Modern telecommunication systems are increasingly built using compact and cost efficient circuits. In particular, the family of low-cost, low-power transceivers has substantially matured in the past two decades. Because of cost issues, high performance semiconductor processes are not normally used for such transceivers. Therefore, high performance is generally achieved through optimum circuit design and innovative techniques.
One of the specifications of a transceiver is the transmitted spectrum phase noise. Often, a transmitted signal is directly or indirectly generated using a local oscillator. Therefore, the transmitted spectrum phase noise performance is tied to the performance of the local oscillator. Phase noise in the local oscillator of a transceiver can overwhelm nearby channels because the phase noise spectral density can grow directly with the transmitted signal power and, at a certain threshold, the phase noise in the signal generated by the local oscillator can be greater than another attenuated signal occupying the same frequency.
FIG. 1 is a block diagram illustrating a conventional transmitter 100. The transmitter 100 consists of a local oscillator 102, a divider 104, a mixer 106, a power amplifier 108, and an antenna 110. The oscillator 102 is a voltage controlled oscillator (VCO). The oscillator 102 is connected to the divider 104 which, in turn, is connected to the mixer 106. The mixer 106 is connected to the power amplifier 108 which, in turn, is connected to the antenna 110.
In operation, after the oscillator 102 generates a signal, various non-linear operations are applied to the generated signal. For example, the signal generated by the oscillator 102 is divided by the divider 104. The divided signal is then mixed with an outside signal 107 by the mixer 106. The mixed signal is then amplified by the amplifier 108 before being transmitted out via the antenna 110.
In some low-end applications, the transmitter 100 is implemented with the oscillator 102 connected directly to the antenna 110. However, in most typical applications, the divider 104, the mixer 106 and the power amplifier 108 are present. In addition, any number of linear buffers or non-linear buffers can be connected between the operational blocks. After the oscillator 102, however, each operational block can add to the noise profile of the signal generated by the oscillator 102, even if the operational blocks following the oscillator 102 are ideally noiseless.
FIG. 2 is a graphical depiction showing a typical phase noise curve of a conventional signal source. The phase noise curve 200 is drawn according to a logarithmic scale and, therefore, the 1/f3 region 202 and the 1/f2 region 204 appear linear with −30 dB/dec and −20 dB/dec slopes, respectively. Depending on the type of the conventional signal source, the 1/f3 region 202 may be substantially large or negligibly small. Due to subsequent buffers or a resistance from non-linear operators immediately after the signal source output, the phase noise profile curve 200 flattens to a minimum thermal noise floor level 206.
For example, a resistor can be coupled to an output of a signal source. The noise from the resistor propagates through the non-linear function of the signal source and increases the noise profile of the signal source. Referring to FIG. 2, the thermal noise floor 206 extends up into a number of the harmonics of the generated signal. Every time a signal with a corresponding phase noise profile goes through a non-linear operation (e.g., division, mixing, non-linear amplification, etc.), frequency components are translated. The translation of the frequency components is accomplished through, for example, an offset equal to the frequency of oscillation or its harmonic frequencies.
FIG. 3 is a functional diagram showing a translation of frequency components during a power amplification process 300 utilizing a conventional Class B power amplifier. The graph 302 is representative of a sample oscillating frequency such as, for example, a frequency generated by a VCO. The power amplifier 304 amplifies the signal 302, passes the portion of the oscillation in the positive input half cycles and zeroes the portion of the oscillation in the negative input half cycles. The resultant output of the power amplifier 304 is represented by the graph 306. The voltage gain during the positive half cycle of the power amplifier 304 illustrated in FIG. 3 can be assumed to be equal to one.
Mathematically, the process reflected on FIG. 3 corresponds to multiplying a cosine wave and a square wave in the time domain. In the frequency domain, the process is represented as a convolution of the impulses of a cosine wave and a series of diminishing impulses of a square wave. Referring to FIG. 4, a graphical depiction 400 of a frequency domain convolution of a sine wave and a square wave is shown. FIG. 4 is representative of the power amplification effect of the power amplifier of FIG. 3 in the frequency domain. The input cosine wave 402 is representative of a signal generated by an oscillator, prior to the application of a non-linear operation (e.g., a power amplification) to the signal. The input cosine signal can be characterized by a thermal noise floor level 403. In this case, the non-linear operations consist of a convolution of the oscillation spectrum with a series of evenly spaced impulses. The square wave 404 is also characterized by a noise profile, which is not reflected in FIG. 4 because the square wave 404 is assumed to be an ideal square wave. The square wave 404 is shown with a DC component, a main impulse at a frequency f0 and additional harmonic impulses with declining amplitudes at respective frequencies 2f0, 3f0, 4f0, etc.
As a result of the convolution process, replicas of the oscillation spectrum are generated and added together. Referring to FIGS. 5A and 5B, there are illustrated graphical depictions 502 and 504 showing a convolution of a cosine wave and an impulse at f0 of a square wave and a convolution of a cosine wave and an impulse at 2f0 of a square wave, respectively. Assuming that the thermal noise floor of the oscillation spectrum in graphical depictions 502 and 504 is a relatively wide band, an accumulation of the thermal noise floor occurs because of the folding of the spectrum onto itself. Therefore, the thermal noise at approximately 2f0 will fold down to approximately f0 due to the convolution of the input cosine with the square impulse at f0 (represented by the graphical depiction 502). Similarly, the thermal noise at approximately 3f0 will fold down to approximately f0 due to the convolution of the input cosine with the square impulse at 2f0 (represented by the graphical depiction 504). The thermal noise level close to the oscillation frequency will, therefore, grow due to the non-linear operation. This characteristic is common to non-linear blocks.
In general, an increase in the thermal noise floor of a generated signal (e.g., an oscillator generated signal) is present even if noiseless blocks (e.g., ideal non-linear operators) follow the signal generator. One of the reasons for this is that the generated signal preserves its thermal noise floor characteristic after it has been generated and even preserves its thermal noise floor characteristic throughout any subsequent non-linear operations since the resulting convolution does not eliminate the thermal noise floor profile.
Further limitations and disadvantages of conventional and traditional approaches will become apparent through comparison of such systems with some aspects of the present invention as set forth in the remainder of the present application with reference to the drawings.